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0.8 ghz to 2.7 ghz direct conversion quadrature demodulator ad8347 rev. a information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to change without notice. no license is granted by implication or otherwise under any patent or patent rights of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ? 2005 analog devices, inc. all rights reserved. features functional block diagram integrated rf and baseband agc amplifiers rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 vdt2 qmxo qopp qofs qopn vag c com3 v gin enbl loip com1 vdt1 qain vps3 28 27 26 25 24 23 22 21 20 19 18 17 16 15 1 2 3 4 5 6 7 8 9 10 11 12 13 14 ad8347 phase splitter phase splitter gain control bias det 02675-001 quadrature phase accuracy 1 typ i/q amplitude balance 0.3 db typ third-order intercept (iip3) +11.5 dbm @ min gain noise figure 11 db @ max gain agc range 69.5 db baseband level control circuit low lo drive ?8 dbm adc-compatible i/q outputs single supply 2.7 v to 5.5 v power-down mode 28-lead tssop package applications figure 1. cellular base stations radio links wireless local loop if broadband demodulators rf instrumentation satellite modems general description baseband level detectors are included for use in an agc loop to maintain the output level. the demodulator dc offsets are minimized by an internal loop, whose time constant is controlled by external capacitor values. the offset control can also be overridden by forcing an external voltage at the offset nulling pins. the ad8347 1 is a broadband direct quadrature demodulator with rf and baseband automatic gain control (agc) amplifiers. it is suitable for use in many communications receivers, performing quadrature demodulation directly to baseband frequencies. the input frequency range is 800 mhz to 2.7 ghz. the outputs can be connected directly to popular a-to-d converters such as the ad9201 and ad9283 . the baseband variable gain amplifier outputs are brought off- chip for filtering before final amplification. by inserting a channel selection filter before each output amplifier, high level out-of-channel interferers are eliminated. additional internal circuitry also allows the user to set the dc common-mode level at the baseband outputs. the rf input signal goes through two stages of variable gain amplifiers prior to two gilbert-cell mixers. the lo quadrature phase splitter employs polyphase filters to achieve high quadrature accuracy and amplitude balance over the entire operating frequency range. separate i and q channel variable gain amplifiers follow the baseband outputs of the mixers. the rf and baseband amplifiers together provide 69.5 db of gain control. a precision control circuit sets the linear-in-db rf gain response to the gain control voltage. 1 u.s. patents issued and pending.
ad8347 rev. a | page 2 of 28 table of contents features .............................................................................................. 1 applications....................................................................................... 1 functional block diagram .............................................................. 1 general description ......................................................................... 1 revision history ............................................................................... 2 specifications..................................................................................... 3 absolute maximum ratings............................................................ 5 esd caution.................................................................................. 5 pin configuration and function descriptions............................. 6 typical performance characteristics ............................................. 8 rf amp and demodulator ......................................................... 8 baseband output amplifiers .................................................... 11 rf amp/demod and baseband output amplifiers .............. 12 equivalent circuits..................................................................... 14 theory of operation ...................................................................... 16 rf variable gain amplifiers (vga)........................................ 16 mixers .......................................................................................... 16 baseband variable gain amplifiers ......................................... 16 output amplifiers ...................................................................... 16 lo and phase splitters............................................................... 16 output level detector ............................................................... 17 bias ............................................................................................... 17 applications..................................................................................... 18 basic connections...................................................................... 18 rf input and matching ............................................................. 18 lo drive interface ..................................................................... 18 operating the vga.................................................................... 19 mixer output level and drive capability .............................. 19 operating the vga in agc mode .......................................... 19 changing the agc setpoint ..................................................... 20 baseband amplifiers.................................................................. 20 driving capacitive loads.......................................................... 21 external baseband amplification ............................................ 21 filter design considerations .................................................... 21 dc offset compensation.......................................................... 22 evaluation board ............................................................................ 23 outline dimensions ....................................................................... 26 ordering guide .......................................................................... 26 revision history 10/05rev. 0 to rev. a updated format..................................................................universal change v gin to v vgin ..........................................................universal changes to figure 46...................................................................... 19 changes to figure 48 ..................................................................... 21 changes to figure 49 and figure 50............................................. 22 changes to ordering guide .......................................................... 27 10/01revision 0: initial version
ad8347 rev. a | page 3 of 28 specifications v s = 5 v; t a = 25c; f lo = 1.9 ghz; v vcmo = 1 v; f rf = 1.905 ghz; p lo = ?8 dbm, r load = 10 k, dbm with respect to 50 , unless otherwise noted. table 1. parameter conditions min typ max unit operating conditions lo/rf frequency range 0.8 2.7 ghz lo input level ?10 0 dbm vgin input level 0.2 1.2 v v supply (v s ) 2.7 5.5 v temperature range ?40 +85 c rf amplifier/demodulator from rfip/rfin to imxo and qmxo (imxo/qmxo load > 1 k) agc gain range 69.5 db conversion gain (max) v vgin = 0.2 v (max gain) 39.5 db conversion gain (min) v vgin = 1.2 v (min gain) ?30 db gain linearity v vgin = 0.3 v to 1 v 2 db gain flatness f lo = 0.8 ghz to 2.7 ghz, f bb = 1 mhz +0.7 db p-p input p1 db v vgin = 0.2 v ?30 dbm v vgin = 1.2 v ?2 dbm third-order input intercept (iip3) f rf1 = 1.905 ghz, +11.5 dbm f rf2 = 1.906 ghz, C10 dbm each tone, (min gain) second-order input intercept (iip2) f rf1 = 1.905 ghz, +25.5 dbm f rf2 = 1.906 ghz, ?10 dbm each tone, (min gain) lo leakage (rf) at rfip ?60 dbm lo leakage (mxo) at imxo/qmxo ?42 dbm demodulation bandwidth ?3 db +90 mhz quadrature phase error f rf = 1.9 ghz ?3 1 +3 degree i/q amplitude imbalance f rf = 1.9 ghz +0.3 db noise figure max gain 11 db mixer agc output level see figure 34 24 mv p-p baseband dc offset at imxo/qmxo, max gain (corrected, ref to vref) 2 mv mixer output swing level at which imd3 = 45 dbc r load = 200 65 mv p-p r load = 1 k 65 mv p-p mixer output impedance 3 baseband output amplifier from iain to iopp/iopn and qain to qopp/qopn r load = 10 k gain 30 db bandwidth ?3 db (see figure 22 ) 65 mhz output dc offset (differential) (v iopp C v iopn ) ?200 50 +200 mv common-mode offset (v iopp + v iopn )/2 ? v vcmo ?40 5 +40 mv group delay flatness 0 mhz to 50 mhz +1.8 ns p-p second-order intermod. distortion f in 1 = 5 mhz, f in 2 = 6 mhz, v in 1 = v in 2 = 8 mv p-p ?49 dbc third-order intermod. distortion f in 1 = 5 mhz, f in 2 = 6 mhz, v in 1 = v in 2 = 8 mv p-p ?67 dbc input bias current +2 a input impedance 1||3 m||pf output swing limit (upper) v s ? 1.3 v output swing limit (lower) 0.4 v
ad8347 rev. a | page 4 of 28 parameter conditions min typ max unit control input/outputs vcmo input @ v s = 2.7 v 1 v @ v s = 5 v 0.5 1 2.5 v gain control input bias current vgin <1 a offset input overriding current iofs, qofs 10 a vref output r load = 10 k 0.95 1.00 1.05 v response from rf input to final bb amp imxo and qmxo connected directly to iain and qain, respectively gain @ v vgin = 0.2 v 65.5 69.5 72.5 db gain @ v vgin = 1.2 v ?3 +0.5 +4 db gain slope ?96.5 ?89 ?82.5 db/v gain intercept linear extrapolation back to theoretical value at vgin = 0 88 94 101 db lo/rf input (see figure 30 through figure 33 for more detail) loip input return loss measuring loip loin, ac-coupled to ground with 100 pf. ?4 db measuring through evaluation board ba lun with termination ?9.5 db rfip input return loss rfip input pin ?10 db enable power-up control low = standby 0 0.5 v power-up control high = enabled +v s ? 1 +v s v power-up time time for final bb amps to be within 90% of final amplitude @ v s = 5 v 20 s @ v s = 2.7 v 10 s power-down time time for supply current to be <4 ma @ v s = 5 v 30 s @ v s = 2.7 v 1.5 ms power supplies vps1, vps2, vps3 voltage 2.7 5.5 v current (enabled) @ 5 v 48 64 80 ma current (standby) @ 5 v 400 a current (standby) @ 3.3 v 80 a
ad8347 rev. a | page 5 of 28 absolute maximum ratings table 2. parameter rating stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. supply voltage vps1, vps2, vps3 5.5 v lo and rf input power 10 dbm internal power dissipation 500 mw ja 68c/w maximum junction temperature 150c operating temperature range ?40c to +85c storage temperature range ?65c to +150c lead temperature (soldering 60 sec) 300c esd caution esd (electrostatic discharge) sensitive device. electros tatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge wi thout detection. although this product features proprietary esd protection circuitry, permanent dama ge may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd pr ecautions are recommended to avoid performance degradation or loss of functionality.
ad8347 rev. a | page 6 of 28 pin configuration and fu nction descriptions rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 vdt2 qmxo qopp qofs qopn vag c com3 v gin enbl loip com1 vdt1 qain vps3 28 27 26 25 24 23 22 21 20 19 18 17 16 15 1 2 3 4 5 6 7 8 9 10 11 12 13 14 ad8347 top view (not to scale) 02675-002 figure 2. 28-lead tssop pin configuration table 3. pin function descriptions equiv. circuit pin no. mnemonic description 1, 28 loin, loip a lo input. for optimum performance, these inputs are differentially driven. typical input drive level is equal to ?8 dbm. to improve the match to a 50 source, connect a 200 shunt resistor between loip and loin. a single-ended drive is possible, but slightly increases lo leakage. 2 vps1 positive supply for lo section. deco uple vps1 with 0.1 f and 100 pf capacitors. 3, 4 iopn, iopp b i-channel differential baseband output. typical o utput swing is equal to 760 mv p-p differential in agc mode. the common-mode level on these pins is programmed by the voltage on vcmo. 5 vcmo c baseband amplifier common-mode voltage. the voltag e applied to this pin sets the output common- mode level of the baseband amplifiers. this pin can either be connected to vref (pin 14) or to a reference voltage from another device (typically an adc). 6 iain d i-channel baseband amplifier input. this pin, which has a high input impedance, should be biased to vref (approximately 1 v). if iain is connected directly to imxo, biasing is provided by imxo. if an ac- coupled filter is placed between imxo and iain, this pin can be biased from vref through a 1 k resistor. the gain from iain to the di fferential outputs iopn/iopp is 30 db. 7, 23 com3 ground for biasing and baseband sections. 8, 22 imxo, qmxo b i-channel and q-channel baseband mixer/vga outputs. low impedance outputs with bias levels equal to vref. imxo and qmxo are typically connected to iain and qain, respectively, either directly or through filters. these outputs have a maximum current limit of about 1.5 ma. this allows for a 600 mv p-p swing into a 200 load. this corresponds to an input level of ?40 dbm @ a maximum gain of 39.5 db. at lower output levels, imxo and qmxo can drive a lower load resistance, subject to the same current limit. 9 com2 rf section ground. 10, 11 rfin, rfip e rf input. rfin must be ac-coupled to ground. the rf input signal should be ac-coupled into rfip. for a broadband 50 input impedance, connect a 200 resist or from the signal side of the rfip coupling capacitor to ground. note that rfin and rfip are no t interchangeable differenti al inputs. rfin is the ground reference for the input system. 12 vps2 positive supply for rf section. deco uple vps2 with 0.1 f and 100 pf capacitors. 13, 16 iofs, qofs f i-channel and q-channel offset nulling inputs. to null the dc offset on the i-channel and q-channel mixer outputs (imxo, qmxo), connect a 0.1 f capa citor from these pins to ground. alternately, a forced voltage of approximately 1 v on these pins disables the offset compensation circuit. 14 vref g reference voltage output. this output voltage (1 v) is the main bias level for the device and can be used to externally bias the inpu ts and outputs of the baseband amplifiers. the vref pin should be decoupled with a 0.1 f capacitor to ground. 15 enbl h chip enable input. active high. 17 vgin c gain control input. the voltage on this pin controls the gain on the rf and baseband vgas. the gain control is applied in parallel to all vgas. the gain control voltage range is from 0.2 v to 1.2 v and corresponds to a gain range from +39.5 db to ?30 db . this is the gain to th e output of the baseband vgas (that is, qmxo and imxo). there is an addition al 30 db of gain in the baseband amplifiers. note that the gain control function has a negative sense (that is, increasing control voltage decreases gain). in agc mode, connect this pin directly to vagc.
ad8347 rev. a | page 7 of 28 equiv. circuit pin no. mnemonic description 18, 20 vdt2, vdt1 d detector inputs. these pins are the inputs to the on-board detector. vdt2 and vdt1, which have high input impedances, are normally connected to imxo and qmxo, respectively. 19 vagc i agc output. this pin provides the output voltage from the on-board detector. in agc mode, connect this pin directly to vgin. 21 vps3 positive supply for biasing and baseband sectio ns. decouple vps3 with 0.1 f and 100 pf capacitors. 24 qain d q-channel baseband amplifier input. bias this high input impedance pin to vref (approximately 1 v). if qain is directly connected to qmxo, biasing is pr ovided by qmxo. if an ac-coupled filter is placed between qmxo and qain, this pin can be biased fr om vref through a 1 k resistor. the gain from qain to the qopn/qopp differ ential outputs is 30 db. 25, 26 qopp, qopn b q-channel differential baseband output. typical o utput swing is equal to 760 mv p-p differential. the common-mode level on these pins is pr ogrammed by the voltage on vcmo. 27 com1 lo section ground. rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 qopn com3 vgin enbl loip com1 vps3 ad8347 phase splitter 1 bias cell det 1 vref vref gain control interface det 2 vref vcmo phase splitter 2 vcmo vdt2 qmxo qopp qofs vag c vdt1 qain 02675-003 4 6 13 8 14 21 12 2 15 10 11 17 20 19 18 22 16 24 25 26 27 23 9 7 1 28 5 3 figure 3. block diagram
ad8347 rev. a | page 8 of 28 typical performance characteristics rf amp and demodulator rf frequency (mhz) 800 gain (db) 2400 2600 ?1.0 ?0.5 0 0.5 1.0 1.5 2.0 2.5 3.0 1000 1200 1400 1600 1800 2000 2200 v s = 2.7v, t a = +25c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +85c 02675-016 v vgin (v) ?35 0.2 mixer gain (db) linearity error (db) 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 ?30 ?25 ?20 ?15 ?10 ?5 0 5 10 15 20 25 30 35 40 45 ?12 ?10 ?8 ?6 ?4 ?2 0 2 4 6 8 10 12 14 t a = ?40c t a = +25c t a = +85c t a = +25c t a = +85c t a = ?40c 02675-013 figure 4. gain and linearity error vs. v vgin , v s = 5 v, f lo = 1900 mhz, f bb = 1 mhz figure 7. gain vs. f lo , v vgin = 0.7 v, f bb = 1 mhz rf frequency (mhz) 800 gain (db) 2400 2600 ? 37 ?36 ?35 ?34 ?33 ?32 ?31 ?30 ?29 1000 1200 1400 1600 1800 2000 2200 ?28 ?27 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +25c 02675-017 v vgin (v) ?35 0.2 mixer gain (db) linearity error (db) 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 ?30 ?25 ?20 ?15 ?10 ?5 0 5 10 15 20 25 30 35 40 45 ?10 ?8 ?6 ?4 ?2 0 2 4 6 8 10 12 14 t a = ?40c t a = +25c t a = +25c t a = +85c t a = ?40c t a = +85c 02675-014 figure 8. gain vs. f lo , v vgin = 1.2 v, f bb = 1 mhz figure 5. gain and linearity error vs. v vgin , v s = 2.7 v, f lo = 1900 mhz, f bb = 1 mhz baseband frequency (mhz) 1 gain (db) 100 30 31 32 33 34 35 36 37 38 10 39 40 41 42 v s = 2.7v, t a = +25c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +85c 02675-018 rf frequency (mhz) 800 gain (db) 2400 2600 30 31 32 33 34 35 36 37 38 39 40 1000 1200 1400 1600 1800 2000 2200 v s = 2.7v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +25c v s = 2.7v, t a = +85c v s = 5v, t a = ?40c v s = 5v, t a = +25c 02675-015 figure 6. gain vs. f lo , v vgin = 0.2 v, f bb = 1 mhz figure 9. gain vs. f bb , v vgin = 0.2 v, f lo = 1900 mhz
ad8347 rev. a | page 9 of 28 baseband frequency (mhz) 1 gain (db) 100 ?5 ?4 ?3 ?2 ?1 0 1 2 3 10 4 5 6 7 8 9 10 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +25c 02675-019 rf frequency (mhz) iip3 (dbm) 5 6 12 7 8 9 800 2400 2600 1000 1200 1400 1600 1800 2000 2200 13 14 15 10 11 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 2.7v, t a = +25c v s = 5v, t a = +85c 02675-022 figure 10. gain vs. f bb , v vgin = 0.7 v, f lo = 1900 mhz figure 13. iip3 vs. f lo , v vgin = 1.2 v, f bb = 1 mhz baseband frequency (mhz) 1 gain (db) 100 ?35 ?34 ?33 ?32 ?31 ?30 ?29 ?28 ?27 10 ?26 ?25 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 5v, t a = +85c v s = 2.7v, t a = +25c 02675-020 rf frequency (mhz) iip3 (dbm) ?30 ?28 ?16 ?26 ?24 ?22 800 2400 2600 1000 1200 1400 1600 1800 2000 2200 ?14 ?12 ?10 ?20 ?18 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 2.7v, t a = +25c v s = 5v, t a = +85c 02675-023 figure 14. iip3 vs. f lo , v vgin = 0.2 v, f bb = 1 mhz figure 11. gain vs. f bb , v vgin = 1.2 v, f lo = 1900 mhz baseband frequency (mhz) iip3 (dbm) 10 0 100 510 15 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 11 12 14 13 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 2.7v, t a = +25c v s = 5v, t a = +85c 02675-024 v vgin (v) 0.2 input p1db (dbm) 1.2 ?35 ?30 ?10 ?5 0 1.11.00.90.80.70.60.50.4 0.3 ?25 ?20 ?15 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 2.7v, t a = +25c v s = 5v, t a = +85c 02675-021 figure 12. input 1 db compression point (op1 db) vs. v vgin , f lo = 1900 mhz, f bb = 1 mhz figure 15. iip3 vs. f bb , v vgin = 1.2 v, f lo = 1900 mhz
ad8347 rev. a | page 10 of 28 baseband frequency (mhz) iip3 (dbm) ?34 0 100 510 ?24 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 ?32 ?30 ?26 ?28 ?22 ?20 ?18 ?16 ?14 ?12 ?10 v s = 2.7v, t a = +85c v s = 2.7v, t a = ?40c v s = 5v, t a = +25c v s = 5v, t a = ?40c v s = 2.7v, t a = +25c v s = 5v, t a = +85c 02675-025 v vgin (v) noise figure (db) 0.2 40 10 30 20 50 60 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 70 ?30 ?25 ?20 ?15 ?10 ?5 0 5 10 15 v s = 2.7v v s = 2.7v v s = 5v 0 iip3 v s = 5v 02675-028 figure 19. noise figure and iip3 vs. v vgin , temperature = 25c, f lo = 1900 mhz, f bb = 1 mhz figure 16. iip3 vs. f bb, v vgin = 0.2 v, f lo = 1900 mhz lo input level (dbm) quadrature phase error (degrees) ?20 ?0.5 ?2.0 ?1.0 ?1.5 0 0.5 ?18 ?16 ?14 ?12 ?10 ?8 ?6 ?4 1.0 ?2.5 ?2 1.5 2.0 2.5 lo frequency = 1900mhz lo frequency = 2700mhz lo frequency = 800mhz 0 02675-029 rf frequency (mhz) iip2 (dbm) 800 40 20 25 35 30 45 50 1000 1200 1400 1600 1800 2000 2200 2400 2600 02675-026 figure 20. quadrature error vs. lo power level, temperature = 25c, v vgin = 0.2 v, v s = 5 v figure 17. iip2 vs. f lo, v vgin = 1.2 v, baseband tone1 = 5 mhz, ?10 dbm, baseband tone2 = 6 mhz, ?10 dbm, temperature = 25c, v s = 5 v lo input level (dbm) noise figure (db) ?20 11.0 9.5 10.5 10.0 11.5 12.0 ?18 ?16 ?14 ?12 ?10 ?8 ?6 ?4 12.5 9.0 ?2 13.0 13.5 14.0 1900mhz 2700mhz 800mhz 0 02675-030 lo frequency (mhz) noise figure (db) 800 12.0 10.0 10.5 11.5 11.0 12.5 13.0 1000 1200 1400 1600 1800 2000 2200 2400 2600 v s = 2.7v v s = 5v 02675-027 figure 21. noise figure vs. lo input level, temperature = 25c, v vgin = 0.2 v, v s = 5 v figure 18. noise figure vs. lo frequency (f lo ), temperature = 25c, v vgin = 0.2 v, f bb = 1 mhz
ad8347 rev. a | page 11 of 28 baseband output amplifiers baseband frequency (mhz) 16 1 100 10 gain (db) 18 20 22 24 26 28 30 32 34 t a = +85c, v s = 5v t a = +25c, v s = 2.7v t a = +25c, v s = 5v t a = +85c, v s = 2.7v t a = ?40c, v s = 2.7v t a = ?40c, v s = 5v 02675-031 baseband frequency (mhz) baseband amplifier output ip3 (dbv rms) ?30 ?25 10 ?20 ?15 ?10 1 100 10 15 20 ?5 0 5 t a = +85c, v s = 5v t a = +25c, v s = 2.7v t a = +25c, v s = 5v t a = +85c, v s = 2.7v t a = ?40c, v s = 2.7v t a = ?40c, v s = 5v 02675-033 v vcmo (v) common-mode offset (mv) ?6 ?4 ?2 0 2 0.5 3.5 2.0 4 6 8 1.0 1.5 2.5 3.0 v s = 5v, mean v s = 5v, mean ? v s = 2.7v, mean + v s = 2.7v, mean v s = 2.7v, mean ? v s = 5v, mean + 02675-034 baseband frequency (mhz) ?25 1 100 10 op1 (dbv rms) ?20 ?15 ?10 ?5 0 5 t a = +85c, v s = 5v t a = +25c, v s = 2.7v t a = +25c, v s = 5v t a = +85c, v s = 2.7v t a = ?40c, v s = 2.7v t a = ?40c, v s = 5v 02675-032 figure 25. common-mode output offset voltage vs. v vcmo , temperature = 25c ( = 1 standard deviation) figure 23. op1 vs. f bb , v vcmo = 1 v
ad8347 rev. a | page 12 of 28 rf amp/demod and baseband output amplifiers baseband frequency (mhz) i to q amplitude mismatch (db) ?0.2 ?0.8 ?0.4 ?0.6 0 0.2 0.4 ?1.0 0.6 0.8 1.0 t a = +85c t a = ?40c t a = +25c 02675-038 0 5 10 15 20 25 30 35 40 v vgin (v) voltage gain (db) ?5 5 45 15 0.2 0.8 0.5 55 25 35 0.3 0.4 0.6 0.7 0.9 1.0 1.1 1.2 65 75 t a = +85c, v s = 5v t a = +85c, v s = 2.7v t a = ?40c, v s = 2.7v t a = ?40c, v s = 5v t a = +25c, v s = 2.7v t a = +25c, v s = 5v 02675-035 figure 29. i/q amplitude imbalance vs. f bb , temperature = 25c, v s = 5 v figure 26. voltage gain vs. v vgin , f lo = 1900 mhz, f bb = 1 mhz rf frequency (mhz) 800 return loss (dbm) 2400 2600 ?12 ?10 ? 8 ? 6 ? 4 ? 2 0 1000 1200 1400 1600 1800 2000 2200 rf without termination rf with termination 02675-039 rf frequency (mhz) 800 quadrature phase error (degrees) 2400 2600 ?2.5 ?2.0 ?1.5 ?1.0 ?0.5 0 0.5 1.0 1.5 1000 1200 1400 1600 1800 2000 2200 2.0 2.5 t a = +85c, v s = 5v t a = ?40c, v s = 5v t a = +25c, v s = 5v 02675-036 figure 27. quadrature phase error vs. f lo , v vgin = 0.7 v, v s = 5 v figure 30. return loss of rfip vs. f rf , v vgin = 0.7 v, v s = 5 v baseband frequency (mhz) quadrature phase error (degrees) 0 ?0.5 ?2.0 ?1.0 ?1.5 0 0.5 5 101520253035 1.0 ?2.5 1.5 2.0 2.5 t a = +85c t a = ?40c t a = +25c 02675-037 40 2.7ghz 2.7ghz 800mhz 800mhz with termination without termination 02675-040 figure 28. quadrature phase error vs. f bb , v vgin = 0.7 v, v s = 5 v figure 31. s11 of rfin vs. f rf , v vgin = 0.7 v, v s = 5 v
ad8347 rev. a | page 13 of 28 rf frequency (mhz) 800 return loss (dbm) 2400 2600 ?12 ?10 ?8 ?6 ?4 ?2 0 1000 1200 1400 1600 1800 2000 2200 ?14 lo port without termination lo port with termination 02675-041 rf input power (dbm) ?70 20 5 15 10 25 30 ?60 ?50 ?40 ?30 ?20 ?10 0 10 0 0.80 0.20 0.60 0.40 1.00 1.20 0 t a = ?40c mixer output voltage (mv p-p) agc voltage (v) t a = ?40c t a = +25c t a = +25c t a = +85c t a = +85c 02675-043 figure 32. return loss of loip vs. f lo , v vgin = 0.7 v, v p = 5 v figure 34. agc voltage and mixer output level vs. rf input power, f lo = 1900 mhz, f bb = 1 mhz, v s = 5 v temperature (c) ?40 supply current (ma) 40 50 45 50 55 60 65 70 75 80 85 ?30 ?20 ?10 60 70 80 v p = 5v v p = 3v v p = 5.5v v p = 2.7v 02675-044 0 102030 with termination without termination 2.7ghz 800mhz 2.7ghz 800mhz 02675-042 figure 33. s11 of loin vs. f lo , v vgin = 0.7 v, v s = 5 v figure 35. supply current vs. temperature, v vgin = 0.7 v, v vcmo = 1 v
ad8347 rev. a | page 14 of 28 equivalent circuits vps3 com3 iain qain 02675-007 vps1 loin loip com1 phase splitter continues 02675-004 figure 36. circuit a figure 39. circuit d vps2 com2 rfin rfip 02675-008 vps3 iopp, iopn, qopp, qopn, imxo, qmxo com3 02675-005 figure 37. circuit b figure 40. circuit e vps3 com3 iofs qofs current mirror 02675-009 vps3 com3 vcmo current mirror 02675-006 figure 41. circuit f figure 38. circuit c
ad8347 rev. a | page 15 of 28 vps3 vag c com3 02675-012 vps3 com3 vref 02675-010 figure 44. circuit i figure 42. circuit g vps3 com3 enbl 02675-011 figure 43. circuit h
ad8347 rev. a | page 16 of 28 theory of operation rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 qopn com3 vgin enbl loip com1 vps3 ad8347 phase splitter 1 bias cell det 1 vref vref gain control interface det 2 vref vcmo phase splitter 2 vcmo vdt2 qmxo qopp qofs vagc vdt1 qain 02675-045 4 6 138 14 21 12 2 15 10 11 17 20 19 18 22 16 24 25 26 27 23 9 7 1 28 5 3 figure 45. block diagram the ad8347 is a direct i/q demodulator usable in digital wireless communication systems including cellular, pcs, and digital video receivers. an rf signal in the frequency range of 800 mhz to 2,700 mhz is directly downconverted to the i and q components at baseband using a local oscillator (lo) signal at the same frequency as the rf signal. differential currents are split and fed to the two gilbert-cell mixers through separate cascode stages. mixers two double balanced gilbert-cell mixers, one for each channel, perform the in-phase (i) and quadrature (q) down conversion. each mixer has four cross-connected transistor pairs that are terminated in resistive loads and feed the differential baseband variable gain amplifiers for each channel. the quadrature lo signals drive the bases of the mixer transistors. the rf input signal goes through two stages of variable gain amplifiers before splitting up to reach two gilbert-cell mixers. the mixers are driven by a pair of lo signals which are in quadrature (90 degrees of phase difference). the outputs of the mixers are applied to baseband i-channel and q-channel variable gain amplifiers. the outputs from these baseband variable gain amplifiers are brought out to pins for external filtering. the filter outputs are then applied to a pair of on-chip, fixed gain, baseband amplifiers. these amplifiers gain up the outputs from the external filters to a level compatible with most a-to-d converters. a sum of squares detector is available for use in an automatic gain control (agc) loop to set the output level. the rf and baseband amplifiers provide approximately 69.5 db of gain control range. additional on-chip circuits allow the setting of the dc level at the i-channel and q-channel baseband outputs, as well as nulling the dc offset at each channel. baseband variable gain amplifiers the baseband vgas also use the x-amp approach with npn differential pairs separated by sections of resistive attenuators. the same interpolator controlling the rf amplifiers controls the tail currents of the differential pairs. the outputs of these amplifiers are provided off chip for external filtering. automatic offset nulling minimizes the dc offsets at both i- and q-channels. the common-mode output voltage is set to the same level as the reference voltage (1.0 v) generated in the bias cell, also made available at the vref pin (see figure 45 ). output amplifiers the output amplifiers gain up the signal coming back from each of the external filters to a level compatible with most high speed a-to- d converters. these amplifiers are based on an active feedback design to achieve high gain bandwidth with low distortion. rf variable gain amplifiers (vga) these amplifiers use the patented x-amp? approach with npn differential pairs separated by sections of resistive attenuators. the gain control is achieved through a gaussian interpolator where the control voltage sets the tail currents supplied to the various differential pairs according to the gain desired. in the first amplifier, the combined output currents from the transconductance cells go through a cascode stage to resistive loads with inductive peaking. in the second amplifier, the lo and phase splitters the incoming lo signal is applied to a polyphase phase splitter to generate the lo signals for the i-channel and q-channel mixers. the polyphase phase splitters are rc networks connected in a cyclical manner to achieve gain balance and phase quadrature. the wide operating frequency range of these phase splitters is achieved by cascading multiple sections of
ad8347 rev. a | page 17 of 28 these networks with staggered rc constants. each branch goes through a buffer to make up for the loss and high frequency roll-off. the output from the buffers then goes into another polyphase phase splitter to enhance the accuracy of phase quadrature. each lo signal is buffered again to drive the mixers. output level detector to create an agc voltage (vagc), two signals proportional to the square of each output channel are summed together and compared to a built-in threshold. the inputs to this rms detector are referenced to vref. bias an accurate reference circuit generates the reference currents used by the different sections. the reference circuit is controlled by an external power-up (enbl) logic signal that, when set low, puts the whole chip into a sleep mode typically requiring less than 400 a of supply current. the reference voltage (vref) of 1.0 v, that serves as the common-mode reference for the baseband circuits, is made available for external use. the vref pin should be decoupled with a 0.1 f capacitor to ground.
ad8347 rev. a | page 18 of 28 applications basic connections rf input and matching the basic connections for operating the ad8347 are shown in the rf input signal should be ac-coupled into the rfip pin and rfin should be ac-coupled to ground. to improve broadband matching to a 50 source, a 200 resistor can be connected from the signal side of the rfip coupling capacitor to ground. figure 46 . the device is powered through three power supply pins: vps1, vps2, and vps3. these pins supply current to different parts of the overall circuit. vps1 and vps2 power the local oscillator (lo) and rf sections, respectively, while vps3 powers the baseband amplifiers. connect all of these pins to the same supply voltage; however, separately decouple each pin using two capacitors. 100 pf and 0.1 f capacitors are recommended, though values close to these can be used. lo drive interface for optimum performance, the lo inputs, loin and loip, should be driven differentially; the m/a-com balun, etc1-1-13 is recommended. unless an ac-coupled transformer is used to generate the differential lo, the inputs must be ac-coupled, as shown in use a supply voltage in the range 2.7 v to 5.5 v. the quiescent current is 64 ma when operating from a 5 v supply. by pulling the enbl pin low, the device goes into its power-down mode. the power-down current is 400 a when operating on a 5 v supply and 80 a on a 2.7 v supply. figure 46 . to improve broadband matching to a 50 source, connect a 200 shunt resistor between loip and loin. a lo drive level of ?8 dbm is recommended. figure 20 shows the relationship between lo drive level, lo frequency, and quadrature error for a typical device. like the supply pins, the individual sections of the circuit are separately grounded. com1, com2, and com3 provide ground for the lo, rf, and baseband sections, respectively. connect all of these pins to the same low impedance ground. figure 47 a single-ended drive is also possible as shown in , but this slightly increases lo leakage. apply the lo signal through a coupling capacitor to loip, and ac-couple loin to ground. because the inputs are fully differential, the drive orientation can be reversed. as in the case of the differential drive, a 200 resistor connected across loip and loin improves the match to a 50 source. rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 qopn com3 vgin enbl loip com1 vps3 ad8347 phase splitter 1 bias cell det 1 vref vref gain control interface det 2 vref vcmo phase splitter 2 vcmo vdt2 qmxo qopp qofs vag c vdt1 qain 760mv p-p differential (agc mode) v cm = 1v qopp qopn lo input ?8dbm 0.8ghz?2.7ghz t1 etc 1-1-13 (m/a-com) 15 3 4 c4 100pf r17 200 c3 100pf 760mv p-p differential (agc mode) v cm = 1v c10 100pf c9 0.1 f c8 100pf c7 0.1 f c6 0.1 f c5 100pf +v s (2.7v?5.5v) rf input 0.8ghz?2.7ghz 0dbm max (agc mode) c1 100pf c2 100pf r1 200 c14 0.1 f 24mv p-p (agc mode) 1v bias (vref) 24mv p-p (agc mode) 1v bias (vref) c13 0.1 f iopp iopn c15 0.1 f 02675-046 4 6 13 8 14 21 12 2 15 10 11 17 20 19 18 22 16 24 25 26 27 23 9 7 1 28 5 3 c16 0.1 f figure 46. basic connections
ad8347 rev. a | page 19 of 28 loin loip ad8347 100pf 100pf 200 lo 02675-047 figure 47. single-ended lo drive operating the vga a three-stage vga sets the gain in the rf section. two of the three stages come before the mixer while the third amplifies the mixer output. all three stages are driven in parallel. the gain range of the first rf vga and that of the second rf vga combined with the mixer are both ?13 db to +10 db. the gain range of the baseband vga is ?4 db to +19.5 db. therefore, the overall gain range from the rf input to the imxo and qmxo pins is ?30 db to approximately +39.5 db. the gain of the vga is set by the voltage on the vgin pin, which is a high impedance input. the gain control function (which is linear-in-db) and linearity are shown in figure 4 and figure 5 at 1.9 ghz. note that the sense of the gain control voltage is negative because as the gain control voltage ranges from 0.2 v to 1.2 v, the gain decreases from +39.5 db to ?30 db. mixer output level and drive capability i- and q-channel baseband outputs, imxo and qmxo, are low impedance outputs (r out @ 3 ) with bias levels equal to v vref , the voltage on pin 14. the achievable output levels on imxo/qmxo are limited by their current drive capability of 1.5 ma maximum. this allows for a 600 mv p-p swing into a 200 load. at lower output levels, imxo and qmxo can drive smaller load resistances, subject to the same current limit. these output stages are not, however, designed to directly drive 50 loads. operating the vga in agc mode although the vga can be driven by an external source such as a dac, the ad8347 has an on-board sum of squares detector to allow the ad8347 to operate in an automatic leveling mode. due to the nature of the detector, an input signal with a higher peak-to-average ratio causes the agc loop to settle with a higher mixer output peak-to-peak voltage. in this data sheet, peak-to-peak calculations assume a sine wave input when referencing agc operation. the connections for operating in this mode are shown in figure 46 . the two mixer outputs are connected to detector input vdt1 and detector input vdt2. the summed detector output drives an internal integrator which, in turn, delivers a gain correction voltage to the vagc pin. a 0.1 f capacitor from vagc to ground sets the dominant pole of the integrator circuit. vagc, which should be connected to vgin, adjusts gain until an internal threshold is reached. this threshold corresponds to a level at the imxo and qmxo pins of approxi- mately 8.5 mv rms. this level changes slightly as a function of rf input power (see figure 34 ). for a cw (sine wave) input, this corresponds to approximately 24 mv p-p. if this signal is applied directly to the subsequent baseband amplifier stage, the final baseband output is 760 mv p-p differential. see the baseband amplifiers section. if the vga gain is set from an external source, vdt1 and vdt2 (the on-board detector inputs) are not used and are tied to vref.
ad8347 rev. a | page 20 of 28 rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loin iain com2 qopn com3 enbl loip com1 vps3 ad8347 phase splitter 1 bias cell det 1 vref vref gain control interface det 2 vref vcmo phase splitter 2 vcmo vdt2 qmxo qopp qofs vag c vdt1 qain qopp qopn lo input ?8dbm 0.8ghz?2.7ghz t1 etc 1-1-13 (m/a-com) 15 3 4 c4 100pf r17 200 c3 100pf 3.8v p-p differential v cm = 2.5v c10 100pf c9 0.1 f c8 100pf c7 0.1 f c6 0.1 f c5 100pf rf input c2 100pf c1 100pf r1 200 c14 0.1 f 120mv p-p 1v bias 120mv p-p 1v bias c13 0.1 f iopp iopn r20 4k r19 1k 2.5v 3.8v p-p differential v cm = 2.5v r21 4k r22 1k +v s +5v 02675-048 4 6 13 8 14 21 12 2 15 10 11 17 20 19 18 22 16 24 25 26 27 23 9 7 1 28 5 3 c16 0.1 f vgin figure 48. adjusting agc level to increase baseband amplifier output swing changing the agc setpoint the agc circuit can be easily set up to level at voltages higher than the nominal 24 mv p-p, as shown in figure 48 . the voltages on pin imxo and pin qmxo are attenuated before being applied to the detector inputs. in the example shown, an attenuation factor of 0.2 (?14 db) between imxo and qmxo and the detector inputs causes the vga to level at approximately 120 mv p-p (note that the resistor divider network must be referenced to v vref ). this results in a peak-to-peak output swing at the baseband amplifier outputs of 3.8 v differential, that is, 1.6 v to 3.4 v on each side. note that v vcmo has been increased to 2.5 v to avoid signal clipping at the baseband outputs. due to the attenuation between the mixer output and the detector input, the variation in the settled mixer output level vs. rf input power will be greater than the variation shown in figure 34 . the variation will be greater by a factor equal to the inverse of the attenuation factor. baseband amplifiers the final baseband amplifier stage takes the signals from imxo and qmxo and amplifies them by 30 db, or a factor of 31.6. this results in a maximum system gain of 69.5 db. when the vga is in agc mode, the baseband i and q outputs (iopn, iopp, qopn, and qopp) deliver a differential voltage of approximately 760 mv p-p (380 mv p-p on each side). the single-ended input signal to the baseband amplifiers is applied at iain and qain, the high impedance inputs. as shown in figure 46 , the baseband amplifier operates internally as a differential amplifier, with the second input driven by v vref . therefore, bias the input signal to the baseband amplifier at v vref . the output common-mode level of the baseband amplifiers is set by the voltage on pin 5, vcmo. connect this pin to vref (pin 14) or to an external reference voltage from a device such as an analog-to-digital converter (adc). v vcmo has a nominal range from 0.5 v to 2.5 v. however, since the baseband amplifiers can only swing down to 0.4 v, higher values of v vcmo are gener- ally required to avoid low end signal clipping. alternatively, the positive swing at each output is limited to 1.3 v below the supply voltage; therefore, the maximum p-p swing is given by 2 (v ps ? 1.3 ? 0.4) v differentially. for example, for the baseband output amplifier to deliver an output swing of 2 v p-p (1 v p-p on each side), v vcmo must be in a range from 0.9 v to 2.5 v.
ad8347 rev. a | page 21 of 28 the differential output offset voltages of the baseband amplifiers are typically 50 mv. this offset voltage results from both input and output effects. filter design considerations baseband low-pass or band-pass filtering can be conveniently performed between the mixer outputs (imxo and qmxo) and the input to the baseband amplifiers. because the output impedance of the mixer is low (approximately 3 ) and the input impedance of the baseband amplifier is high, it is not practical to design a filter that is reactively matched to these impedances. an lc filter can be matched by placing a series resistor at the mixer output and a shunt resistor (terminated to v vref ) at the input to the baseband amplifier. the overall signal-to-noise ratio can be improved by increasing the vga gain by driving it with an external voltage or by changing the setpoint of the agc circuit. see the changing the agc setpoint section. driving capacitive loads in applications where the baseband amplifiers are driving unbalanced capacitive loads, place some series resistance between the amplifier and the capacitive load. for example, for a 10 pf load, use four 200 series resistors, one in each baseband output. because the mixer output drive level is limited to a maximum current of 1.5 ma, the characteristic impedance of the filter should be greater than 50 , especially to achieve larger signal swings. external baseband amplification reduce baseband output offset voltage and noise by bypassing the internal baseband amplifiers and amplifying the mixer output signal using a high quality differential amplifier. in the example shown in figure 50 shows the schematic for a 100 , fourth-order elliptic low-pass filter with a 3 db cutoff frequency of 20 mhz. source and load impedances of approximately 100 ensure that the filter sees a matched source and load. this also ensures that the mixer output is driving an overall load of 200 . note that the shunt termination resistor is tied to vref and not to ground. the frequency response and group delay of this filter are shown in figure 49 , two ad8132 differential amplifiers are used to gain up the mixer output signals by 20 db. in this example, the setpoint of the agc circuit was increased to give an approximate 72 mv p-p input to the external amplifiers. this resulted in final baseband output signals of 720 mv p-p. figure 51 and figure 52 . the closed-loop bandwidth of the amplifiers in figure 49 is equal to approximately 20 mhz. higher bandwidths are achievable, but at the cost of lower closed-loop gain. in imxo ad8347 vref vdt1 (see text) l 3 1.2 h r3 2 iain r4 2 c1 4.7pf c3 8.2pf c2 150pf rs 95.3 l 1 0.68 h c4 82pf rl 100 02675-050 c16 0.1 f figure 49 , the output common-mode levels at pin 2 (v ocm pin) of the ad8132s are set by the ad8347s vref (approximately 1 v). the output common- mode levels can also be externally set, using, for example, the reference voltage from an adc. imxo ad8347 vref 10 f 0.1 f qmxo vdt1 720mv p-p differential v cm = 1v r19a 4.99k +5v 10 f 0.1 f ad8132 r17a 499 r18a 499 r22 20k r23 10k 72mv p-p vdt2 72mv p-p r25 20k r24 10k r17b 499 4.99k r20a 4.99k r19b +5v ?5v r18b 499 720mv p-p differential v cm = 1v ?5v 4.99k r20b 10 f 0.1 f 10 f 0.1 f ad8132 02675-049 3 8 2 1 6 4 5 3 8 2 1 6 4 5 c16 0.1 f frequency (mhz) ?80 1 100 10 attentuation (db) ?70 ?60 ?50 ?40 ?30 ?20 ?10 0 02675-051 figure 49. external baseband amplification example figure 51. frequency response of 20 mhz baseband low-pass filter
ad8347 rev. a | page 22 of 28 frequency (mhz) 0 1 100 10 group delay (ns) 5 10 15 20 25 30 35 40 45 50 02675-052 dc offset compensation feedthrough of the lo signal to the rf input port results in self-mixing of the lo signal. this produces a dc component at the mixer output that is frequency dependent. the ad8347 includes an internal circuit that actively nulls any dc offsets that appear at the mixer output. the dc bias level of the mixer output (which should ideally equal v vref , the bias level for the baseband sections of the chip) is continually com- pared to v vref . any differences between the mixer output level and v vref forces a compensating voltage on to the mixer output. the time constant of this correction loop is set by the capacitors that are connected to pin iofs and pin qofs (each output can be separately compensated). for normal operation, 0.1 f capacitors are recommended. the corner frequency of the compensation loop is given approximately by figure 52. group delay of 20 mhz baseband low-pass filter if the vga is operating in agc mode, the detector inputs (vdt1 and vdt2) can be tied either to the inputs or outputs of the filter. connecting the detector inputs to the inputs of the filter (imxo and qmxo) causes the vga leveling point to be determined by the composite of the wanted signal and any unfiltered components, such as blockers or signal harmonics. alternatively, connecting vdt1 and vdt2 to the outputs of the filters ensures that the leveling point of the agc circuit is based upon the amplitude of the filtered output only. the latter option is more desirable as it results in a more constant baseband output. however, when using this method, set the leveling point of the agc so that the out-of-band blockers do not overdrive the mixer output. () ofs ofs db c c f = the corner frequency must be set to a frequency that is much lower than the symbol rate of the demodulated data. this prevents the compensation loop from falsely interpreting the data stream as a changing offset voltage. to disable the offset compensation circuits, tie iofs and qofs to vref.
ad8347 rev. a | page 23 of 28 evaluation board figure 53 shows the schematic of the ad8347 evaluation board. note that uninstalled components are indicated with the open designation. the board is powered by a single supply in the range of 2.7 v to 5.5 v. table 4 details the various configuration options of the evaluation board. rfin vref rfip vps2 imxo com3 iopp iofs iopn vcmo vps1 loip iain com2 vdt2 qmxo qopp qofs qopn vag c com3 enbl loin com1 vdt1 qain vps3 28 27 26 25 24 23 22 21 20 19 18 17 16 15 1 2 3 4 5 6 7 8 9 10 11 12 13 14 ad8347 1 5 3 4 r17 200 c3 100pf c2 100pf t1 etc 1-1-13 j3 lo c5 100pf c6 0.1 f tp1 +v s j6 iopn r35 0 j5 iopp r36 0 j11 vcmo c1 0.1 f l3 (open) l2 (open) l1 (open) c18 (open) c4 (open) c19 (open) c22 (open) c20 (open) c21 (open) c17 (open) l4 (open) l6 (open) l5 (open) c30 (open) c26 (open) c31 (open) c25 (open) c29 (open) c28 (open) c27 (open) +v s j1 qopn r37 0 j2 qopp r38 0 j8 qmxo j9 vagc j10 vgin r34 (open) r33 0 lk4 lk6 tp5 c9 0.1 f c10 100pf r40 (open) lk3 c15 0.1 f c14 0.1 f tp6 a b v pos sw1 lk2 c16 0.1 f tp3 tp2 +v s lk1 j7 imxo j4 rfip c13 0.1 f c8 100pf c7 0.1 f r18 200 c12 100pf c11 100pf r39 (open) lk5 r8 (open) r6 0 02675-053 vgin tp4 figure 53. evaluation board schematic
ad8347 rev. a | page 24 of 28 02675-054 figure 54. silkscreen of component side 02675-056 02675-055 figure 55. layout of component side figure 56. layout of circuit side
ad8347 rev. a | page 25 of 28 table 4. evaluation board configuration options component function default condition tp1, tp4, tp5 power supply and ground vector pins. not applicable tp2, tp6 iofs and qofs probe points. not applicable tp3 vref probe point. not applicable lk1, j11 baseband amplifier output bias. installi ng this link connects vref to vcmo setting the bias level on the baseband amplifiers to vref, which is equal to approximately 1 v. alternatively, the bias level of the baseband amplifiers can be set by applying an external voltage to sma connector j11. lk1 installed lk2, lk6, lk3, j9, j10 agc mode. installing lk2 and lk6 connects imxo and qmxo, the mixer outputs, to vdt2 and vdt1, the detector inputs. by installing lk3, which connects vgin to vagc, the agc mode is activated. the agc voltage can be observed on sma connector j9. with lk3 removed, apply the gain control signal for the internal variable gain amplifiers to sma connector j10. lk2, lk6, lk3 installed lk4, lk5, j7, j8 r6, r33, l1 to l5 c4, c17 to c22, c25 to c31 r8, r34, r39, r40 baseband filtering. installing lk4 and lk5 connects imxo and qmxo, the mixer outputs, directly to iain and qain, the ba seband amplifier inputs. with r6 and r33 installed (0 ), iain and qain can be observed on sma connector j7 and sma connector j8. by removing lk4 and lk5 and installing r8 and r34, lc filters can be inserted between the mixer outputs and th e baseband amplifier inputs. r8 and r34 can be used to increase the effective o utput impedance of imxo and qmxo (these outputs have low output impedances). r 39 and r40 can be used to provide terminations for the filter at iain and qa in (high impedance inputs.) terminate r39 and r40 to vref. lk4, lk5 installed r6 = r33 = 0 (size 0603) l1 to l5 = open (size 0805), c4, c17 to c22, c25 to c31 = open (size 0805), r8 = r34 = open (size 0603), r39 = r40 = open (size 0603) r35, r36, r37, r38 baseband amplifier output series resistors. r35 = r36 = r37 = r38 = 0 (size 0603) sw1 device enable. when in position a, the enbl pin is connected to +v s and the ad8347 is in operating mode. in position b, the enbl pin is grounded, putting the device in power-down mode. sw1 = a
ad8347 rev. a | page 26 of 28 outline dimensions compliant to jedec standards mo-153-ae 28 15 14 1 8 0 seating plane c oplanarit y 0.10 1.20 max 6.40 bsc 0.65 bsc pin 1 0.30 0.19 0.20 0.09 4.50 4.40 4.30 0.75 0.60 0.45 9.80 9.70 9.60 0.15 0.05 figure 57. 28-lead thin shrink small outline package [tssop] (ru-28) dimensions shown in millimeters ordering guide model temperature range package description package option ad8347aru ?40c to +85c 28-lead tssop ru-28 ad8347aru-reel7 ?40c to +85c 28-lead tssop, 7 tape and reel ru-28 ad8347aruz ?40c to +85c 28-lead tssop ru-28 1 ad8347aruz-reel7 ?40c to +85c 28-lead tssop, 7 tape and reel ru-28 1 AD8347-EVAL evaluation board 1 z = pb-free part.
ad8347 rev. a | page 27 of 28 notes
ad8347 rev. a | page 28 of 28 notes ? 2005 analog devices, inc. all rights reserved. trademarks and registered trademarks are the property of their respective owners. c02675-0-10/05(a)

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